This invention relates to the field of switching DC-to-DC power conversion and in particular to the new class of switching converters employing two novel methods: lossless switching method and method for novel magnetics structure. Lossless switching improves substantially the conversion efficiency, while new DC Transformer structure either minimizes or entirely eliminates the DC energy storage in magnetics core structures hence resulting in a very compact size of magnetics and efficiency improvements. Additional performance benefits are in increased DC overload current capability and reduced EMI noise with improved reliability.
The following notation is consistently used throughout this text in order to facilitate easier delineation between various quantities:
1. DCxe2x80x94Shorthand notation historically referring to Direct Current but by now has acquired wider meaning and refers generically to circuits with DC quantities;
2. ACxe2x80x94Shorthand notation historically referring to Alternating Current but by now has acquired wider meaning and refers to all Alternating electrical quantities (current and voltage);
3. i1, v2xe2x80x94The instantaneous time domain quantities are marked with lower case letters, such as i1 and v2 for current and voltage;
4. I1, V2xe2x80x94The DC components of the instantaneous periodic time domain quantities are designated with corresponding capital letters, such as I1 and V2;
5. xcex94i1xe2x80x94The difference between instantaneous and DC components is designated with xcex94, hence xcex94i1 designates the ripple component or AC component of current i1;
6. iCCxe2x80x94The composite current equal to sum of currents through the input switch S1 and complementary input switch Sxe2x80x21, that is iCC=iS1+iSxe2x80x21 
The following common defining relationships and notations related to magnetic circuit descriptions is used consistently throughout:
1. Flux linkage xcex is the total flux linking all N turns and is defined as xcex=N"PHgr" and "PHgr" is the total flux in the magnetic core;
2. Inductance L is defined as the slope of the xcex-i characteristic, i.e., L=xcex/i;
3. Flux density B is the flux per unit area defined by B="PHgr"/S where S is a magnetic core cross-section area;
Present invention imposes also a need to introduce completely new terminology for the two major novelties, neither of which is present in prior-art switching converter terminology:
1. New magnetic devices with substantially reduced size and increased efficiency made possible by a special switching converter structure and corresponding magnetic circuit structure;
2. Novel methods of switching, which make possible the complete elimination of switching losses (except for gate-drive losses) and thus result in the highest possible efficiency improvement.
The new magnetic devices come in two basic variants named as follows:
1. DC Transformer is a special magnetic structures with multiplicity of inductor windings on a common magnetic core, in which DC current flow of each winding and AC voltage polarity of each winding as imposed by a present invention non-isolated switching converter are such to result in the reduction of the total DC ampere-turns of all winding and hence in reduced DC flux in the common magnetic core and in some instances even substantially zero total DC Ampere-turns and substantially zero DC flux in the common core.
2. Isolated DC Transformer is a special magnetic structures having inductors and isolation transformer windings with the same performance features as DC Transformer but having in addition a galvanic isolation between the source and load.
Lossless Switching Methods require new definition of the switches, switching intervals and transition intervals they create as well as the respective duty ratio D as follows:
1. S1, S2, Sxe2x80x21, Sxe2x80x22xe2x80x94Switch designations respectively for input switch, output switch, complementary input switch, and complementary output switch and, at the same time, designate the switching states of the respective active, controllable switches as follows: high level indicates that active switch is turned-ON, low (zero) level that active switch is turned-OFF;
2. Dxe2x80x94The duty ratio is defined as D=tON/TS where tON is the ON time interval during which the input switch is closed (turned ON) and TS is the switching period defined as TS=1/fS where fS is a switching frequency;
3. Dxe2x80x2xe2x80x94The complementary duty ratio Dxe2x80x2 is defined as Dxe2x80x2=tOFF/TS where tOFF is the OFF time interval during which the input switch S1 is open (turned OFF);
4. State-1 Intervalxe2x80x94The time interval during which input switch S1 and output switch S2 are turned-ON (closed), while complementary input switch Sxe2x80x21 and complementary output switch Sxe2x80x22 are both turned OFF (open);
State-2 Intervalxe2x80x94The time interval during which input switch S1 and output switch S2 are both turned OFF (open), while complementary input switch Sxe2x80x21 and complementary output switch Sxe2x80x22 are both turned ON (closed);
6. (1-2) transition intervalxe2x80x94The time interval between State-1 and State-2 interval during which, in precisely defined sequence and timing, input switch S1 and output switch S2 reverse their state from ON to OFF while complementary input switch Sxe2x80x21 and complementary output switch Sxe2x80x22 reverse their state from OFF to ON;
7. (2-1) transition intervalxe2x80x94The time interval between State-2 and State-1 interval during which, in precisely defined sequence and timing, input switch S1 and output switch S2 reverse their state from OFF to ON while complementary input switch Sxe2x80x21 and complementary output switch Sxe2x80x22 reverse their state from ON to OFF;
8. CR2, CRxe2x80x22xe2x80x94Designation for the output switch and complementary output switch implemented as a current rectifier (CR) diodes and their corresponding switching time diagram. Since diode is a two-terminal passive switch, switching time diagram represents also the state of diode switch as follows: high level indicates that the diode is ON and low level that diode is OFF;
9. Ixe2x80x94Designates one quadrant switch operating in the first quadrant. The Roman number (I through IV) within a rectangular box around ideal switch signifies limitation to particular one-quadrant operation;
10. CBSxe2x80x94Together with the rectangular box around the ideal switch and this symbol designates the Current Bi-directional Switch (CBS) as a three-terminal, controllable semiconductor switching device, which can conduct the current in either direction when turned ON, but blocks the voltage of only one polarity when turned OFF.
Switching Converter Categorizations
Over the last two decades a large number of switching DC-to-DC converters had been invented with the main objective to improve conversion efficiency and reduce the converter size. The past attempts to meet both of these objectives simultaneously have been hampered by the two main problems, which up to now seemed to be inherent to all switching DC-to-DC converters:
1. The large DC current bias present in the filtering inductors at either input or output of the converters (as well as the DC-bias current present in the isolation transformer of some of the isolated converters) resulted in a big size of the magnetic components, since an air-gap proportional to the DC current bias must be inserted in the AC flux path in order to prevent magnetic core saturation. This also resulted in a very inefficient use of the magnetic material, which was largely wasted. Even a relatively small air-gap, in the order of 1 mm (40 mils), drastically reduces the total inductance. This loss of inductance was compensated by either an inordinately large increase of the switching frequency (hence increase of losses) or by increasing the size of the magnetic cores, or both.
2. An implementation of soft switching methods to reduce significant switching losses at increased switching frequencies was DC load current dependent and required for its operation an unwanted large output inductor AC current ripple (larger than twice the magnitude of the maximum DC load current) thereby diminishing most of the recovered energy due to increased conduction losses caused by this large AC ripple current. Other soft-switching methods also resulted in additional undesirable loss mechanisms.
Magnetic Circuit Categorizations
The past efforts to solve the first problem and reduce the large size and weight of the magnetic components, inductor and transformer, together with the new method of the present invention has resulted in three major categories relative to the implementation of the magnetic components:
1. Separate Magnetics category in which all magnetic components are used as separate magnetic devices, one or more inductors each with DC current bias, and isolation transformer with or without DC current bias. This realization leads to maximum size and weight of the magnetic components and large energy storage due to the DC bias current in individual magnetic components.
2. Coupled-inductors and Integrated Magnetics category in which magnetic components are combined into a single magnetic structure in which total DC energy storage remained substantially equal to the DC energy storage of separate magnetic components when summed together. This magnetic realization resulted in size and weight reduction and some efficiency improvement, but the major problem associated with DC energy storage remained.
This clearly motivated the search for solution in a form of a switching converter of present invention with novel magnetic structure.
3. DC transformer category in which magnetic components are combined into a single magnetic structure in such a way that the total DC energy storage is reduced and in some cases completely eliminated.
Switching Method Classification
The demand for reduced size and weight of electronic power processing equipment to make it compatible with ever shrinking size of electronic signal processing equipment resulted in the continuous push toward increasing the switching frequency at which DC-to-DC switching converters operate: from initial 20 kHz level to 200 kHz and higher switching frequencies. This, in turn, results in proportionally increased switching power losses. Hence, the past efforts to reduce the converter conduction and switching losses resulted in a number of switching methods, which together with a new lossless switching method of the present invention constitute three broad categories:
1. Hard-switching method in which no attempts were made to reduce the switching losses;
2. Soft-switching method in which measures were taken to reduce the switch losses. Unfortunately, in most cases, the reduction of switching losses was accompanied by the increase of other losses, such as, for example, conduction losses of the switching devices due to the requirement for increased AC ripple currents or losses associated with energy stored in transformer leakage inductance or an additional external resonant inductor. These byproduct power losses clearly led to smaller net loss reduction and modest efficiency improvements. Thus, a third method relative to the switching mechanism is needed and is introduced in a present invention as:
3. Lossless switching method which fully eliminates the extra losses (except for gate-drive losses) of the soft-switching method and thereby results in substantial efficiency improvement.
Categorization by Number of Switches
The switching converters can also be classified into three major converter classes relative to a number of power switches employed, such as:
1. Two-Switch Converter class, example of which is the prior-art buck converter.
2. Three-Switch Converter class such as prior-art forward converter;
3. Four-Switch Converter class such as prior-art forward, half-bridge, and push-pull switching DC-to-DC converters.
The present invention employs four switches and thus belongs to the Four-Switch Converter class.
Prior-art Problems with DC Current Bias and Magnetic Core Saturation
The problem associated with the DC-bias of magnetic components (inductors and transformers) can be best understood with reference to the classical prior-art buck converter shown in FIG. 1a and the accompanied output inductor current waveform of FIG. 1b. Since the converter output supplies DC power to the load, the inductor in the buck converter must pass the DC component of the load current, which is IDC. Hence, it clearly cannot be designated as an ordinary inductor used in alternating current (AC) applications as the inductor in FIG. 2a. An AC inductor is wound on magnetic core material in order to substantially increase its inductance value. For example, typical ferrite core material has at room temperature a relative permeability on the order of xcexcr=3000. Hence the inductance of the coil is magnified by a factor of 3000 simply by inserting the magnetic core material without any air gap as in FIG. 2a. The corresponding flux linkage xcex versus current i characteristics is as in FIG. 2b with a high slope illustrating the high inductance value L (maximum attainable with that core material). The flux linkage excursions caused by the AC current are symmetrical around the center of the magnetic core operating characteristic (FIG. 2b). Even if a very small DC current IDC shown in FIG. 2b were to pass through the coil, the magnetic core material would saturate and instead of the desirable large inductive impedance, the inductor would look like a short circuit. Thus, to avoid core saturation, all present switching converters xe2x80x9csolvexe2x80x9d this DC-bias problem in a xe2x80x9cbrute-forcexe2x80x9d way by inserting an air-gap in the magnetic flux path as illustrated in FIG. 3a. This clearly reduces the inductance value proportionally to the inserted air-gap size (the larger the DC current, the bigger air-gap is needed, and hence the smaller is the resulting inductance value), as seen by the flux linkage characteristic of FIG. 3b for an un-gapped core (dotted lines) and gapped core (full lines) and their corresponding inductances L and Lg. Clearly three very detrimental factors did occur:
1. By insertion of the air-gap, the inductance value is drastically reduced. It is not uncommon to see the original un-gapped inductance L reduced by a factor of xcex1=100 to xcex1=1000 with the air-gap included. In order to compensate for this reduction of inductance, the switching frequency should be increased or a bigger core size is used, or combination of both.
2. The already small available AC flux linkage excursions due to the low saturation flux density BSAT of 0.3 T (telsa) for ferrite material, is further reduced due to the presence of the DC-bias in the core. For example, in typical applications, the DC-bias might correspond to a flux density of 0.25 T thus leaving only 0.05 T for the superimposed AC flux density. AC flux density excursions are indicated in thick lines in FIG. 3b. To increase the AC flux density excursions, larger core size is required or increased switching frequency, or both.
3. Due to the presence of positive DC bias, only one part of the saturation characteristic is utilized and another part of xcex94B=BSAT=0.3 T is wasted.
The DC-bias problem is not only limited to all inductors used up to now in DC-to-DC converters but is also present in many isolation transformers, such as for example in the popular flyback converter shown in FIG. 4a. This transformer does provide galvanic isolation and the ability to step-up or step-down the voltage through the transformer turns ratio, but contrary to the ordinary AC line transformer, it has a large DC-bias and requires a correspondingly big air-gap as shown in FIG. 4b. Hence the magnetic core is biased in one direction thus limiting the superimposed AC flux excursions as seen in FIG. 4c. 
Let us now also quantify DC-bias effect on an output inductor design for a 5 V, 100 W buck converter. For a DC load current of I2=20 A, and number of winding turns N=6 implemented on a ferrite core with saturation flux density BSAT=0.3 T (telsa), BDC=0.2 T is available for the DC-bias and the remaining 0.1 T for the superimposed AC flux excursions. To support NI=120 ampere-turns the required air-gap is calculated from (lg=xcexc0NI/BDC=30 mils=0.75 mm), where (xcexc0=4xcfx8010xe2x88x927 H/m) is the permeability of free space. If L is the inductance without air-gap, and Lg is the inductance with air-gap lg=0.75 mm, then the ratio of the two inductances is given by L/Lg=xcexcr lg/lm=50 where xcexcr is the relative permeability of the ferrite material, which for typical materials used in switching converters xcexcr=3000, and lm=45 mm is the mean magnetic path length of the core used. Thus, the maximum available inductance of a given core is reduced by a factor of 50. At higher power and especially higher DC load current levels this becomes progressively much more severe. It is not uncommon for some high power DC converter applications in the kilowatt range to see that after ferromagnetic material was inserted the inductance increased only by a factor of 2 or 3 over the inductance without any magnetic material due to the large air-gap needed to prevent saturation. Clearly, this is a tremendous waste of the magnetic material, which has the ability to increase the inductance 3000 times over that of an air-core coil. This is also the reason, why in switching converters in which isolation transformer has no DC bias, such as in the isolated Cuk converter, the transformer size is several times smaller in size and weight in comparison with the size and weight needed for the input and output inductors, which by far dominate the size and weight of the switching converter and also result in increased losses.
The loss of the inductance due to insertion of the air-gap in the flux path is compensated either by increasing the core cross-section or by increasing the switching frequency, or a combination of both would rapidly degrade the overall efficiency. Thus, it is very desirable to either reduce the DC-bias in the magnetic core, or, if possible, to eliminate it entirely.
In the past, there had been a number of attempts to correct this fundamental limitation of DC-to-DC converters, but with a very limited success. One approach was followed by magnetic manufacturers, such as Hitachi and TDK. In the article xe2x80x9cReducing Magnetic Component Size with Reverse Biased Ferrite Corexe2x80x9d published in the Proceedings of the Powercon 6 conference, May 1979, author Shiraki (of Hitachi) proposed to add a permanent magnet to the air-gap and hence by proper orientation of the permanent magnet produce reverse bias the in the core in the direction opposite to the DC-bias created by the current of the magnetics winding as shown in FIG. 5a. The net effect is that the AC flux excursions are now extended into the negative core flux swing area as seen in FIG. 5b and FIG. 5c and would allow the core cross-section and volume reduction by up to 50%. The TDK corporation developed a line of PCH cores based on their reverse biased core modification as reported in the Proceedings of Powercon 9, July 1982 in article xe2x80x9cA New Reverse Biased Choke Coilxe2x80x9d by Nakamura and Ohta of TDK corporation. Note, however, that both approaches also include an air-gap and operate along the reduced, xe2x80x9cthick linkxe2x80x9d slope as shown in FIG. 5b and FIG. 5c. Hence, the large reduction of inductance from its maximum inductance capability of the un-gapped core (dashed line in FIG. 5b and FIG. 5c) is still present. Another drawback is that the core can only support the designed-in maximum DC-ampere-turns. If this is exceeded, the core will saturate and the overload capability will be lost. Since the permanent magnet provides a fixed reverse bias independent of the DC load current, at no-load current, the magnetic flux is entirely along the negative part of the core flux saturation characteristic (FIG. 5c). In fact, the permanent magnet generates the maximum allowable bias but in the negative (reverse) direction. This will be compared later with the novel DC Transformer of the present invention in which there is an automatic self-balancing, such that at any DC load current, total DC Ampere-turns of all windings is zero. The other practical limitations, such as increased cost of the special cores with inserted permanent magnets, the extra loss due to added core loss of the permanent magnet, etc. rendered this approach unattractive, which is by now abandoned by both of these companies.
Another attempt to reduce or eliminate the DC-bias problem is to make use of a special converter circuit configuration instead of a special magnetic core structure. Such an approach is disclosed in U.S. Pat. No. 5,166,869 issued to Bryce L. Hesterman for xe2x80x9cComplementary Electronic Power Converterxe2x80x9d in which a xe2x80x9ccomplementary transformerxe2x80x9d is introduced. This transformer combines the input and output inductors into a coupled-inductor configuration in which the DC flux generated by the input inductor DC current is canceled by the flux generated by the output inductor DC current. The main drawback of the proposed converter is that it is capable of producing only the fixed input to output voltage conversion ratio determined by a fixed turns ratio of the two windings. Hence it cannot provide a regulated voltage through pulse-width modulation of the switches even over a limited input voltage range. From another point of view, there are other fixed conversion ratio converters such as 50% driven bridge type converters, which do not need inductors with DC-bias current for either input or output filtering, hence the DC-bias problem is not present. Thus, a highly desirable objective is to have a switching converter with a variable conversion ratio, capable of handling a wide range of input voltages and provide regulated output, and at the same time either completely eliminate the DC-bias or reduce it substantially.
Another possible approach is to combine input and output inductor windings into a common coupled-inductor structure as shown in FIG. 6a and as was disclosed in U.S. Pat. No. 4,184,197, xe2x80x9cDC-to-DC Switching Converterxe2x80x9d by S. Cuk and R. D. Middlebrook and U.S. Pat. No. 4,257,087, xe2x80x9cDC-to-DC Switching Converter with Zero Input and Output Current Ripple and Integrated Magnetics Circuitsxe2x80x9d by S. Cuk. As described in the above patents, the basic prerequisite for combining the two windings on a common core is to have identical AC voltages across the two inductors before the coupling, and that the AC voltage matching is maintained over a wide operating range of duty ratio D as illustrated by the identical AC voltage waveforms in FIG. 6b (duty ratios D1 and D2) for the converter of FIG. 6a. In practical applications, a small mismatch of the AC voltages could be absorbed gracefully due to the ever-present leakage inductance between the two windings as explained below. Since the AC voltages are identical, the placement of the two windings on the same core in a coupled-inductor structure imposes the requirement for equal number of turns N (AC voltage ratio equal to turns ratio as in an ideal transformer), because in the simplified model the leakage inductance is not included. The proper understanding of the AC voltage polarity marking in coupled-inductor and integrated magnetic structures (polarity markings with dot-marked ends as in FIG. 6a) and the actual directions of the instantaneous and DC currents relative to those dot markings (currents i1 and i2 and their DC components I1 and I2 in FIG. 6a) is of critical importance for understanding not only previous inventions but is crucial for understanding the present invention.
Note the difference of this coupled-inductor structure and a transformer. The output inductor instantaneous current i2 in the coupled-inductors of FIG. 6a flows into the dot-marked end, whereas in an AC transformer, the secondary current i2 flows out of the dot-marked terminal. Thus, the corresponding DC component I2 of the load current in the coupled-inductor structure also flows into the dot-marked end. Consequently, the generated DC fluxes "PHgr"1 and "PHgr"2 add together (FIG. 7c) resulting in a combined flux vs. ampere-turns characteristic of FIG. 7f. 
Clearly, the air-gaps g1 and g2 of the two corresponding separate inductors of FIG. 7a and FIG. 7b add, resulting in larger total air-gap g1+g2 for the coupled-inductor core structure of FIG. 7c. Note that due to the larger total air-gap, the total effective permeance P in FIG. 7f (and hence corresponding inductance) is still further reduced from permeances of the separate cores in FIG. 7d and FIG. 7e. 
The main advantage of the coupled-inductor structure is that it can reduce the ripple current on the output side dramatically and even produce zero output ripple current, as first disclosed in U.S. Pat. No. 4,184,197. As disclosed in U.S. Pat. No. 5,790,005 xe2x80x9cLow Profile Coupled Inductors and Integrated Magneticsxe2x80x9d, the inventors E. Santi and S. Cuk have shown that the air-gap position plays the key role in zero ripple current adjustment. When the air-gap is solely placed on the side of input inductor as in FIG. 8a, the total leakage inductance LL effectively appears solely on the output inductor side as in the model of FIG. 8b. Since the converter of FIG. 6a generates identical AC voltages on the input and output inductors, the net AC voltage across this leakage inductance is zero (xcex94v=vL1xe2x88x92vL2=0) leading to zero ripple current (xcex94i2=0) in the output inductor.
Note that the ripple current on the input inductor remains relatively large due to presence of the air-gap. The only way to reduce that ripple would be to reduce the air-gap. Thus, one might be tempted to connect on purpose the coupled-inductors of FIG. 7c into the converter of FIG. 6a so that the output inductor dot-marked end is reversed and connected as in FIG. 9a to the junction between diode CR1 and capacitor C1. Note that with such connection the output inductor DC current I2 will flow out of the dot-marked end. Hence, at least for one duty ratio D=0.5, and provided equal number of turns are used on both windings, a complete DC flux cancellation could be accomplished in the coupled-inductors magnetic core. Thus, the air-gap could be eliminated since the DC-ampere-turns of the two windings cancel. However, elimination of the ripple current is not possible even for this single operating point, since the model in FIG. 9b clearly points out that the small residual leakage inductor would now be subject to an AC voltage, which is two times larger than the input inductor AC voltage vL1 resulting in huge circulating ripple current for both input and output inductors.
Clearly, what is needed is a special switching converter which inherently has the opposing flow of the DC currents in the input and output inductor windings (into the dot-marked end and out of dot-marked end respectively) and yet the respective AC voltage waveforms at the two inductors windings should be in phase with each other at respective dot-marked ends. Further constraint is to have identical or closely matching magnitudes of both AC voltages and DC currents. Yet an additional constraint is to maintain the above relationship over a wide operating range, that is a wide change of the duty ratio D. Note that even the first constraint of opposing DC current flows (for the net DC-ampere-turn reduction, if not complete cancellation) and the in-phase waveforms of the respective AC voltages is not realized in the prior art converter of FIG. 6a as well as in all other Coupled-inductors and Integrated Magnetics structures proposed in the past.
Out of a large number of possible switching converters, with input and output inductors only a handful of them even meet the pre-requisite for coupling them on a common magnetic core, that is to have identical AC voltage waveforms, both in terms of their in-phase relationship as well as their magnitudes. Thus, imposing the additional even more severe constraints, such as opposing DC current flows as well as their matching magnitudes, may appear at first too restrictive and impossible to achieve at all. This, however, is not the case, and solution is found in the form of the DC Transformer realization presented in Summary Section and Section on Detailed operation.
Prior-Art Problems with Hard-Switching and Soft-Switching Converters
One of the first soft-switching methods which provided reduction of switching losses was introduced by C. Henze, H. C. Martin and D. W. Parsley in xe2x80x9cZero-Voltage Switching in High-Frequency Power Converters Using Pulse-Width Modulationxe2x80x9d, IEEE Applied Power Electronics Conference, (IEEE Publication 88CH2504-9) pp33-40, 1988 record on a basic buck converter which belongs to Two-Switch Converter class and is shown in prior-art of FIGS. 10(a-e). In order to obtain zero-voltage switching at a constant switching frequency, the usual transistor-diode implementation of two switches is replaced with two MOSFET transistors, each of which is modeled as a parallel connection of an ideal switch with an anti-parallel parasitic body-diode and a parasitic drain-to-source capacitance, resulting in circuit models of FIGS. 10(a-d). The total switching cycle TS is broken down into 4 intervals by proper drive timing of the two switches S and Sxe2x80x2 as shown in FIG. 10e. Note that with two controllable switches, two well defined transition intervals are introduced during which both switches are OFF. The first transition interval (tN in FIG. 10e), starts when switch S is turned OFF (as in FIG. 10a) and is also known as the xe2x80x9cnaturalxe2x80x9d transition. By turning OFF the switch S, the inductor current IP is flowing naturally in a needed direction (represented by the current source IP on FIGS. 10a-d). This current source IP charges the parasitic capacitance CS of switch S and discharges parasitic capacitance CSxe2x80x2 of switch Sxe2x80x2 until capacitance CSxe2x80x2 is fully discharged at which instant the body-diode of switch Sxe2x80x2 clamps the voltage at zero and prevents reverse charging of parasitic capacitance CSxe2x80x2 of switch Sxe2x80x2. At that instance, the switch Sxe2x80x2 can be turned ON with zero switching losses (FIG. 10b), since the charge of CSxe2x80x2 was already relocated to capacitance CS of the switch S (charged to Vg). Hence the converter state as in FIG. 10c is obtained for interval Dxe2x80x2TS. In order to perform the reverse process during the second transition interval, a negative inductor current lN is needed. The simplest method to accomplish this is to design the output inductor to have a large ripple current, such that its peak-to-peak ripple current is more than 2 times the maximum DC load current. As seen in the inductor current waveform in FIG. 10e, the instantaneous inductor current iL will at some point during Dxe2x80x2TS interval reverse direction and become negative with magnitude IN. Just before the end of complementary interval Dxe2x80x2TS the switch Sxe2x80x2 is turned OFF initiating the so-called xe2x80x9cforcedxe2x80x9d transition (since the inductor current is now intentionally forced to become negative by the converter circuit designed for large ripple). During this forced transition interval (lF in FIG. 10e), the converter states of FIGS. 10(c-d) apply and opposite to tN interval occurs: this negative inductor current IN charges parasitic capacitance CSxe2x80x2 of switch Sxe2x80x2 and discharges parasitic capacitance CS of switch S until voltage VS of S reaches zero. At that instant body-diode clamps the voltage on switch S to zero forcing switch S to turn-ON at zero voltage in a lossless manner. Hence recycling of the charge stored in the parasitic capacitances CS and CSxe2x80x2 is provided instead of being dissipated each cycle as in xe2x80x9chard-switchingxe2x80x9d.
Even though soft-switching can be achieved on both active switches S and Sxe2x80x2 in this very simple manner, and the voltage stresses on the switches are as low as in the original hard-switching buck converter, the big disadvantage is that the magnitude of the output inductor ripple current must be higher than two times the maximum DC load current and this must be satisfied for a wide range of the operating duty ratio D, which makes output inductor ripple current requirement even higher. This, in turn, increases the conduction losses significantly and diminishes savings obtained by reduced switching losses. In addition, an increased size of output capacitor is needed to absorb this large ripple current and to reduce the output AC ripple voltage to acceptable level.
Another prior-art method of reduction of switching losses belongs to the Three-Switch Converter class, as disclosed by U.S. Pat. No. 4,415,959 issued to P. Vinciarelli, for xe2x80x9cForward Converter Switching at Zero Currentxe2x80x9d. To force the main input power switch to switch at zero current in this quasi-resonant converter, the reactive components, small resonant inductor and small resonant capacitor are used to distort the main switch square-wave like current waveform into a sinusoidal-like current waveform. This makes possible turning ON and OFF of the main switch at zero current and reduces the switching losses caused by switch current and switch voltage overlap and by finite switching time characteristic of the semiconductor switching devices. Unfortunately, the increased RMS value of the switch current increases the conduction losses, thereby diminishing some of the switching loss reduction gained by zero current switching. More serious, however, is the fact that the dominant switching loss due to xc2xdCV2 energy stored on the parasitic capacitance of the main switch still remains and is dissipated when that switch is turned ON. This switching loss is especially pronounced in applications operating from high input DC voltages, such as nominal 300 V DC input voltage in OFF-line applications, using rectified AC line as a DC source.
The converter disclosed in U.S. Pat. No. 4,441,146 issued to P. Vinciarelli for xe2x80x9cOptimal resetting of the transformer""s core in single-ended forward convertersxe2x80x9d belongs to the Four-Switch Converter class. The branch comprising the auxiliary switch and storage capacitance, and placed on transformer secondary was used with a sole purpose to form a xe2x80x9cmagnetizing current mirrorxe2x80x9d to reset the transformer""s magnetic core and has not other roles. On the contrary, in the present invention, the branch comprising an auxiliary switch and an auxiliary capacitor is placed on the primary side of the novel switching converter topology accomplishing not only the transformer""s magnetic core reset but also more importantly the elimination of switching losses.
The converter disclosed in the U.S. Pat. No. 5,291,382 issued to Isaac Cohen for xe2x80x9cPulse Width Modulated DC/DC Converter With Reduced Ripple Current Component Stress and Zero Voltage Switching Capabilityxe2x80x9d also belongs to the Four-Switch Converter class. In this converter, the soft-switching at zero voltage is achieved in a method analogous to the buck converter of FIGS. 10(a-e). It is based on the small magnetizing inductance of the isolation transformer which results in large magnetizing ripple current, hence with the same soft-switching and efficiency limitations as in a soft-switching buck converter. However, since soft-switching is accomplished by large magnetizing ripple current of transformer and not with a large output inductor ripple current as in a buck converter, the undesirable effect of large output inductor ripple current of the buck converter on output ripple voltage is eliminated.
Yet another example of the Four-Switch Converter class is the prior-art converter disclosed in the U.S. Pat. No. 5,066,900 issued to John Basset for xe2x80x9cDC/DC Converter Switching at Zero Voltagexe2x80x9d. In this converter, the leakage inductance of the transformer is used as a resonant inductor to force the reduction of switching losses. However, the use of the passive rectifier diodes for the two switches on the converter""s output (secondary side) instead of the controllable switches with optimum switching time control as in the present invention, severely limits the loss reduction which can be achieved with this soft-switching technique.
The common to all above cited prior-art soft-switching converters is that although they employ different soft-switching methods on the members of Three-switch and Four-switch Converter class, they all utilize only the passive current rectifier switches for the two output switches. Even when synchronous rectifiers are implemented, their switching coincides with that of the replaced rectifier diodes, resulting in similar switching loss characteristics. To the contrary, the present invention, which belongs to the Four-switch Converter class uses in addition to the two active switches on the input sides also two active and controllable switches on the output secondary side, which are the Current Bidirectional Switch (CBS) semiconductor devices. Together with a special switching sequence and time control of all four controllable switching devices, present invention results in reduction of switching losses without increase of other losses, such as conduction losses, leakage losses, etc., as was the case with the prior-art soft-switching methods.
Present invention introduces novel lossless switching methods, which require specific converter topology, proper type of semiconductor switches, and precise sequence and drive timing for the four controllable switches.
A primary objective of this invention is to provide a switching DC-to-DC converter with the following three basic features:
1. Ultra high efficiency;
2. Elimination of the switching losses enabling very high switching frequency and reduction of the converter size and weight;
3. Very small size of the magnetics with ultra high output DC overload current capability.
All these basic features can be simultaneously realized with the present invention due to its novel characteristics:
1. New switching converter topology with four controllable switches;
2. Novel lossless switching method in a novel lossless switching topology;
3. New DC transformer structure with further reduction of size and method for near elimination of the stored DC energy in magnetics with further increase of efficiency.
Switching Converter Topology with Four Controllable Switches
The new lossless switching DC-to-DC converter is comprised of a Power Processing Stage with four controllable MOSFET switches and the Switching Time Control Box, which provides the needed switching sequence and timing control for all controllable switches to achieve lossless switching in a number of alternative ways. The invention is embodied in either non-isolated converter or isolated converter.
The isolated Power Processing Stage is comprised of an isolation transformer, input and output inductors, series input capacitor, auxiliary capacitor and four controllable switches. The input inductor is connected in series with the DC source and provides the non-pulsating (continuous) input current, while the output inductor is connected in series with the DC load and provides non-pulsating (continuous) output current. The input capacitor is connected in series with the input inductor and transformer primary. Input switch and complementary input switch are on the transformer""s primary side, while output switch and complementary output switch are on the transformer""s secondary side. The branch with the complementary input switch in series with auxiliary capacitance is positioned within converter in such a way to conduct AC ripple current only while the complementary input switch is closed.
Lossless Switching Method and Lossless Switching Topology
The AC ripple current of the complementary input switch together with the controllable output switch and the novel switching sequence and time control enables lossless switching operation, with efficiency and size performance not possible with prior-art soft-switching converters. The lossless switching method utilizes a resonance between the parasitic capacitances of the primary side switches and the leakage inductance of the isolation transformer in a novel way obtained through precise sequence and timing of switching of all controllable switches. Such special sequence and timing control results in the resonant current with three components, which provides switching with significantly reduced losses in comparison to classical soft-switching method, hence designated lossless switching.
DC Transformer Structure and Method for Reduction of Stored DC Energy
In further improvement of the present invention, the input inductor, the isolation transformer and the output inductor are combined on a common magnetic core to form a new magnetic device, an Isolated DC Transformer, with unique features. Conventional magnetic structures have large DC flux and thus need to include an air-gap to prevent saturation of the magnetic core with consequent loss of inductance and increase of overall size. In the novel Isolated DC Transformer, however, the combined DC-ampere-turns of all windings cancel, resulting in zero DC flux and hence elimination of air-gaps in the magnetic cores with consequent increase of inductances and decrease in size. Thus, the Isolated DC Transformer without an air-gap has high DC output current overload capability, small size and weight, and provides desirable ripple-free DC input and DC load currents. The DC stored energy is also reduced to zero leading to corresponding increase in efficiency.